Enhanced digital communications receiver using channel impulse estimates

ABSTRACT

The present invention provides for a receiver that employs integrated symbol timing adjustment and carrier tracking algorithms based on channel impulse response estimates. It uses variable coefficient least mean square tracking during estimation of the transmission channel&#39;s impulse response. Maximum likelihood sequence estimation equalization can also be integrated into the system using a non-real time operation that permits and exploits &#34;time-reversed&#34; equalization to significantly enhance bit-error rate performance and permits implementation of all the functions in a reasonably simple receiver.

This application is a division of U.S. Pat. application Ser. No.07/722,440, filed Jun. 27, 1991, U.S. Pat. No. 5,263,026, assigned tothe same assignee as the present invention.

BACKGROUND

The present invention relates generally to digital cellularcommunications, and more particularly, to a maximum likelihood sequenceestimation based equalization method for use in mobile digital cellularreceivers.

Communication channels in the cellular environment commonly impose acombination of distorting effects on transmitted signals. Rayleighfading, where a signal's perceived power level rises and falls rapidlyover a wide range, results from the combination of signals that havetraversed paths differing in length by at least a significant fractionof a wavelength (i.e., about 30 cm. for cellular). Differences in pathtransmission times that approach the time taken to transmit a symbolresult in a second problem called delay spread.

Delay spread results in reception of multiple delayed replicas of atransmitted signal. Each Rayleigh faded replica has randomly distributedamplitude and phase, and the rate at which this complex quantity variesis constrained by the Doppler bandwidth associated with a vehicle'sspeed. In a frequency nonselective environment, the sampled outputs of areceiver's matched falter provides uncorrelated estimates of thetransmitted data. As such, in terms of discrete time samples, thechannel has exhibited an impulse response proportional to a deltafunction. With delay spread, on the other hand, the discrete timechannel impulse response is extended to introduce energy at a number ofsymbol times. The effect of the channel on the transmitted signal, inturn, may be viewed as the convolution of the transmitted informationwith the channel's impulse response. The channel, therefore, emulates aconvolutional coding process.

This leads to the possibility of estimating the transmitted informationthrough the use of methods analogous to typical decoding ofconvolutional codes, i.e., maximum likelihood sequence estimationtechniques. Unlike the more widely applied forward error correctiondecoding environment, the details of the encoding process in a reverseerror correction decoding environment, are not known a priori by thereceiver. Issues related to the need to estimate the form of theencoding process are addressed by this invention.

For the North American digital cellular system, a number of documentsdefine the standards of implemented components. With respect to thisinvention, the following are of interest: "Dual-Mode Mobile Station -Base Station Compatibility Standard" denoted here as IS-54, [EIA/TIAProject Number 2398, Rev. A, Jan. 1991]and "Recommended MinimumPerformance Standards for 800 MHz Dual-Mode Mobile Stations", denotedhere as IS-55, [EIA/TIA Project Number 2216, Apr. 1991].

It would therefore be desirable to provide an enhancement to theprocessing performed by equalizers for use in mobile telephones thatprovides for system complexity reduction and that provides for betterperformance in a fading channel.

SUMMARY OF THE INVENTION

The present invention provides for an equalization method that forms thenucleus of a receiver for digital cellular mobile telephones. A numberof distinct aspects of the design are considered novel, including:non-real time operation mode that permits and exploits "time-reversed"equalization, which significantly enhances bit error rate performance;the use of maximum likelihood sequence estimation; use of variablecoefficient least mean square tracking during estimation of thetransmission channel's impulse response; and integrated symbol timingadjustment and carrier tracking algorithms.

More particularly, the present invention comprises a method ofprocessing received symbols including known preamble data andtransmitted data, which method compensates for the effects of a powerfade caused by a frequency selective fading channel. One specific aspectof the present invention comprises a method of processing samplesreceived from a delay-spread, fading channel, which samples areassociated with a block of data transmitted within a time slot, andwherein the method is adapted to make data decisions, and comprises thefollowing steps: (1) Storing samples received during the time slot; (2)Estimating the location within the time slot at which decision errorsare most probable, by determining the location in time of the maximumfade depth in the transmission channel impulse response; (3) Processingthe stored samples, starting with the first received sample andproceeding beyond the location of the maximum fade depth, using apredetermined maximum likelihood sequence estimation procedure togenerate estimates of the transmitted data; (4) Processing the storedsamples, starting with the final received sample and proceeding in areverse direction with respect to the time sequence in which the sampleswere stored, beyond the abovedetermined location, using the maximumlikelihood sequence estimation process to generate estimates of thetransmitted data; (5) Simultaneous with the above two processing steps,generating estimates of the characteristics of the transmission channelimpulse response, which are used in the maximum likelihood sequenceestimation process; (6) Processing the output of the preceding estimategenerating steps to generate data decisions.

The step of generating the channel impulse response estimates typicallycomprises using variable tap coefficients that are determined byestimating tap settings within the estimated channel impulse response byminimizing the square of the difference between actual received samplesand those synthesized by passing known transmitted signals through theestimated channel. The processing is done in an iterative manner bycombining previous estimates of channel impulse response and newestimates thereof based on recent information, and by varying the ratioof the contributions from the previous and new estimates as a functionof location within the time slot.

The method may further comprise a symbol timing adjustment procedurecomprising the following steps: (1) Using a subset of the samplesreceived during a time slot, generating an error measurement comprisinga measure of the degree to which the estimated channel impulse responsematches the actual channel impulse response; (2) Generating a pluralityof similar measures utilizing simultaneously recorded samples havingdifferent time offsets wherein at least one sample is advanced and onesample is retarded in time relative to the above measure; (3) Searchingfor a bit time setting that minimizes the above error measurement andadjusting the sampling to reflect the newly determined bit time setting.

The method may also further comprise a carrier offset tracking methodcomprising the following steps: (1) Recording at least two samples of atleast one tap within the estimated channel impulse response at selectedsymbol locations within the time slot during the equalization process;(2) Generating frequency offset estimates at each of a plurality of timeslots by observing the phase difference in each time slot between the atleast two samples; (3) Combining this plurality of frequency offsetestimates by using a filtering process to generate a precise frequencyoffset estimate; (4) Adjusting a controllable frequency source tocompensate for the precise frequency offset estimate.

Also, the present invention also uses a forward maximum likelihoodestimation method of processing samples received from a delay-spread,fading channel, which is adapted to make data decisions. This methodcomprises the following steps: Storing samples received during the timeslot; Processing the stored samples, starting with the first receivedsample and proceeding beyond the last received sample of the time slotusing a predetermined maximum likelihood sequence estimation procedureto generate estimates of the transmitted data; Simultaneous with theabove processing, generating estimates of the characteristics of thetransmission channel impulse response which are used in the maximumlikelihood sequence estimation process; Processing the outputs of thepreceding steps to generate data decisions.

The maximum likelihood sequence estimation method offers significantperformance advantages when compared with alternative equalizationoptions, such as decision feedback equalization, for example. Withrespect to meeting industry defined standards for performance of digitalcellular telephones used in fading environments, the performance of theequalizer incorporating the maximum likelihood sequence estimationprocess of the present invention stands out as superior. It is the onlyknown process that is capable of meeting, or even approaching, thecurrent digital cellular mobile telephone specifications. Time reversedoperation further enhances performance and permits implementation withreasonable complexity of a standards compliant mobile receiver.

BRIEF DESCRIPTION OF THE DRAWINGS

The various features and advantages of the present invention may be morereadily understood with reference to the following detailed descriptiontaken in conjunction with the accompanying drawings, wherein likereference numerals designate like structural elements, and in which:

FIG. 1 illustrates the problem associated with reception in a mobileenvironment with a fading channel and one aspect of the solutionprovided by the present invention;

FIG. 2 is a block diagram of a digital cellular mobile telephonereceiver incorporating a maximum likelihood sequence estimation basedequalizer in accordance with the principles of the present invention;

FIG. 3 shows the processing performed in the maximum likelihood sequenceestimation based equalizer of FIG. 2;

FIG. 4 illustrates a graph of relative error versus bit timing offsetthat illustrates one problem overcome using the maximum likelihoodsequence estimation based equalizer of the present invention;

FIG. 5 is a flow diagram illustrating the processing performed by theequalizer of the present invention to implement carrier frequency offsetcompensation; and

FIGS. 6A and 6B are flow diagrams illustrating the processing performedby the equalizer of the present invention to implement bit timing.

DETAILED DESCRIPTION

Referring to the drawing figures, FIG. 1 illustrates the problemassociated with reception in a mobile environment having a fadingchannel and one aspect of the solution provided by the presentinvention. FIG. 1 shows a graph showing received power level from atypical fading channel versus time. The location of the power fade isshown relative to a typical time slot. The time slot is shown enlargedand includes a preamble and a coded digital verification color codefield (CDVCC), which comprise known data that is used to initialize areceiver system employing the equalizer of the present invention. At thelowest portion of FIG. 1, equalization processing in accordance with thepresent invention is illustrated with the arrows "A" and "B", in whichthe equalizer of the present invention processes forward and timereversed computations through the location of the power fade in order toaccomplish the objectives of the present invention. This will be morefully described below with reference to FIGS. 2 and 3.

FIG. 2 is a block diagram of a digital cellular mobile telephonereceiver system 20 incorporating a maximum likelihood sequenceestimation based equalizer 21 in accordance with the principles of thepresent invention. The system 20 comprises an amplifier 22 whose outputis coupled by way of a downconverter, comprising a frequency source 23and a mixer 24, to an analog filter 25. An analog to digital converter26 is coupled to the analog filter 25 in order to digitize thedownconverted data. A matched filter 27 is coupled between the analog todigital converter 26 and the equalizer 21 of the present invention. Theequalizer 21 comprises a memory 30, a 4-state equalization trellis 31that is adapted to calculate maximum likelihood sequence estimationmetrics, a channel impulse response estimator 32, and an equalizercontrol circuit 33.

A serially coupled AGC circuit 35 and gain control circuit 38 arecoupled to the amplifier 22. The equalizer control circuit 33 is coupledto an output of the matched filters 27 and is coupled to an input to thefrequency source 23. Symbol sampling (bit timing) time control circuitry37 is coupled to the equalizer control circuit 33 and the acquisitioncircuit 36 and provides control signals to the analog to digitalconverter 26. The output of the matched filters 27 is coupled to the AGCcircuit 35 and the acquisition circuit 36 and to the equalizer controlcircuit 33 that is employed to control the frequency source 23 andprovide training data for use in initializing the equalizer 21.

In operation, a partially filtered IF signal with a center frequency of85.05 MHz enters the gain controllable amplifier 22. The resultingsignal is then downconverted using the frequency source 23 and the mixer24 to 461.7 kHz. This signal is then filtered using a narrow analogfilter 25 to reject most of the received signals outside the 30 kHz bandof interest. The resulting signal is then sampled and converted to 8-bitdigital samples using the analog to digital (A/D) converter 26. A 16 tapfractionally spaced digital FIR filter 27 then performs matchedfiltering to produce symbol spaced samples which enter the equalizer 21.Temporally offset matched filters 34 that are substantially the same asthe matched filters 27 are provided for use by the symbol timing controlcircuit 37, via the equalizer control circuit 33.

The principles of maximum likelihood sequence estimation employed in theequalizer 21 have been described in technical literature starting in theearly 1970's. A useful outline is found in "Adaptive Maximum-LikelihoodReceiver for Carrier-Modulated Data Transmission Systems", by G.Ungerboeck, IEEE Trans. on Communications, Vol. COM-22, pp. 624-636, May1974. Another description of the maximum likelihood sequence estimationtechnique is provided in the reference "Digital Communications - 2ndEdition.", by J. G. Proakis, 1989, pp. 610-642.

The maximum likelihood sequence estimation process is outlined asfollows. The channel has an impulse response containing significantenergy in, say, N symbols. Assume that the transmitter sends a sequenceof symbols, much longer than N. The transmitted sequence may bedescribed as the transitions between states, where each statecorresponds to a group of N- 1 transmitted symbols. The states,therefore, correspond to overlapping groups of transmitted symbols. Inconsecutive states, therefore, all but one constituent symbol are thesame, and the possible transitions between states are correspondinglyconstrained. As each sample is received, the equalization trellis 31considers every possible sequence of N symbols that could havecontributed to its value, by convolving that sequence with the estimatedchannel impulse response. For each hypothesized sequence, the result ofthe convolution corresponds, or fails to correspond, in some way(defined by a statistic called a metric) to the measured sample. On anindividual basis, the hypothesized sequence with the closest match tothe measured sample (the best metric) is the most likely to have beentransmitted. However, over many samples and under the constraint thatonly certain state transitions are possible, the path (sequence ofstates) with the minimum cumulative metric has maximum likelihood, andthis is what the decoder selects.

The system 20 has no a priori knowledge of the form of the encoderemployed in the transmitter. Performance of the equalizer 21 thereforedepends on the accuracy of the estimate of the encoder's state, thechannel impulse response (CIR). FIG. 2 also shows the signals used inestimating the channel impulse response. The objective is to estimatethe form of the transversal finite impulse response filter that wouldtake as its input the transmitted information symbols {a(n)}, andproduce at its output the samples taken from the matched filter, {z(n)}.During the transmission of preambles and coded digital verificationcolor codes, the receiver knows the values of {a(n)}. However, at othertimes, only the estimated values {a_(d) (n)} are available for use inthe channel impulse response estimation process. This dependence leadsto a significant perfor- mance-degrading possibility. If decision errorsemerge from the equalizer, and these are then used to update theestimate of the channel impulse response, then further decision errorsbecome more probable leading in a circular fashion to further decisionerrors and breakdown of the equalization process. This phenomena isreferred to as a "channel impulse response tracking breakdown". Suchdifficulties are most likely to arise at the periods of minimumsignal-to-noise ratio, or when the received signal power is at itsminimum during reception of a slot.

Within the IS-54 standard, which describes the interface between mobileand base equipment for North American digital cellular systems, eachinformation time slot is preceded by a known sequence, designated as the"preamble". As viewed by the receiver, therefore, information in thetime slot is bounded on both sides by known sequences; the preamble forthis slot and the preamble for the subsequent slot. Consequently, thisequalizer 21 is adapted to mitigate the effects of a channel impulseresponse tracking breakdown. By finding the most probable instant atwhich the problem might occur, equalizer operation approaches thatinstant from both forward and a time-reversed directions, both of whichbegin with known information sequences that are useful for training.Assuming that a channel impulse response tracking breakdown occurs, thisapproach minimizes the number of affected symbols by predicting thefailure point and avoiding equalization beyond that point.

At 100 km/hr, which is the maximum speed specified in IS-55, whichdescribes the mobile unit minimum performance requirements, the averagetime between fades are on the order of 12 milliseconds. Given time slotdurations of about 6.7 milliseconds, there is only a small possibilityof two significant fades occurring within a time slot. However, veryclose to the center of the slot is the coded digital verification colorcode field. Even after a channel impulse response tracking breakdown,the channel impulse response estimator 32 is very likely to recoverduring processing of the coded digital verification color codes due tothe certainty of the transmitted data. Hence, the underlying period forwhich multiple fades are a concern is around 3.5 milliseconds. Thechance of more than one deep fade occurring during this time is verylow. Consequently, time-reversed equalization improves bit error rateperformance in the digital cellular environment.

The present equalizer 21 uses a 4-state architecture, corresponding toN=2, where N is the length of the estimated channel impulse response.This choice assumes that the energy in two (symbol-spaced) samples ofthe channel's impulse response dominates. To avoid channel impulseresponse tracking breakdown problems, reverse equalization is used forthose symbols following the minimum power point in a received time slot.

More specifically, FIG. 3 shows the processing performed in the maximumlikelihood sequence estimation based equalizer 21 of FIG. 2. The firststep involves finding the location of the power fade (box 51 ) in termsof symbol number. Processing starts in the forward direction toward thelocation of the power fade. The symbol number is set to 0 (box 52), andthen incremented (box 53). A decision is made whether the symbol thenprocessed is a training symbol (box 54). If the symbol encountered is atraining symbol, then training data is inserted (box 57). If a trainingsymbol is not processed, then the equalization trellis is employed togenerate metrics and, if possible, a decision (box 55). This isaccomplished using equations outlined below. Then it is determined if adecision has been made (box 56). If a decision has been made, then anestimate of the channel impulse response is generated (box 58). If thedecision is not made, or once the channel impulse response estimate hasbeen generated, then the symbol number is compared to the location ofthe power fade plus a predetermined number of additional symbols (box59). Processing is then repeated by incrementing the symbol number (box53) and repeating steps (boxes 54-59) until the fade location plus apredetermined number of additional symbols has been reached.

Once the desired symbol location is reached in (box 59), then processingis performed in the reverse direction starting with the preamble of thenext succeeding time slot, namely symbol number 177, for example. Thesymbol number is set to 178 (box 62), and then decremented (box 63). Adecision is made whether the symbol then processed is a training symbol(box 64). If the symbol encountered is a training symbol, then trainingdata is inserted (box 67). If a training symbol is not processed, thenthe equalization trellis is employed to generate branch metrics and adecision (box 65). This is accomplished using the equations outlinedbelow. Then it is determined if a decision has been made (box 66). If adecision has been made, then an estimate of the channel impulse responseis generated (box 68). If the decision is not made, or once the channelimpulse response estimate has been generated, then the symbol number iscompared to the location of the power fade less a predetermined numberof additional symbols (box 69). Processing is then repeated bydecrementing the symbol number (box 63) and repeating steps (boxes64-69) until the fade location less a predetermined number of additionalsymbols has been reached.

More particularly, and in operation, samples entering the equalizer 21may be identified as z(n), and the output decisions may be identified asa(n). The probability of correctness of a(n) depends on location withinthe bursts. When a(n) is known with certainty the values of a(n),denoted at(n), are used by the channel impulse response estimator 32 fortraining. At other times, the best estimate of a(n) is the output of thetraceback decision process of the equalization trellis 31, denoted a_(d)(n).

The equalization trellis 31 operates as follows. Equalization proceedsin the forward direction from the beginning of the preamble up until Msymbols after the minimum power symbol. In the reverse direction, thesame occurs with processing continuing M symbols beyond the minimumpower point. This overlap ensures that trace-back through the trellis inall likelihood converges to a single path by the minimum power point.

Traceback for actual decisions does not occur until the completion ofthe equalization process. In addition to final traceback, however, thereis a need for tentative decisions during equalization, to provide dataestimates for the channel impulse response estimation to remain current.A trade-off in determining these tentative decisions arises (a) becausethe more up-to-date the information is, the more up-to-date the channelimpulse response estimate can be (remembering that the channel is farfrom stationary at high speeds), and (b) the higher the number ofsymbols that are considered before tentative decisions are made, themore accurate the decisions will be; and hence, the lower theprobability that errors are introduced into the channel impulse responseestimation. In the case of 4-state equalization there is very littlesensitivity to the number of constraint lengths of delay introduced.

Branch metrics are calculated in the equalizer 21 using the followingequation: ##EQU1## where app₋₋ state(l) represents a hypothetical statein combination with potential input data; a_(h) (l,n) is a correspondingtransmitted signal (constellation point), C represents the currentestimate of the channel's impulse response, and z is the measured outputof the matched filter 27.

The channel estimator 32 utilizes a second order least mean squarealgorithm to determine the coefficients of the transversal filter 27that is an estimate of the channel. ##EQU2## where C₀ (k) and C₁ (k) arecomplex values of estimated channel impulse response taps, C_(S0) (k)and C_(n) (k) are complex intermediate values related to the estimatedchannel impulse response taps, permitting second order operation, K₁ andK₂ are the real gain values controlling the tracking rate of the channelimpulse response estimation process, z(k) are complex symbol spacedsampled outputs of the receiver matched filter, and a(k) are complexestimated or known values of transmitted symbols.

The values K₁ and K₂ within these equations control the rate ofadaptation, and (conversely) the sensitivity to noise and decisionerrors. Consequently, to minimize the error rate, a trade-off betweenability to track changes in the channel and degradation in performancedue to imperfect input information is needed to optimize the values ofK₁ and K₂. The optimal values of K₁ and K₂ vary as a function ofinstantaneous signal to noise ratios, and thus as a function of depth offade. Therefore, algorithms for modifying the values during each bursthave been evaluated, with considerable improvement in performancerelative to that achievable with constant settings.

One approach for modifying K₁ and K₂ has provided good performance andis as follows:

1. Set the values of K₁ and K₂ that will apply aat the symbol determinedto correspond to the deepest fade; K₁₋₋ fade.

2. Adjust each value linearly (with preset slope - K₁₋₋ slope and K₂₋₋slope) to reach the selected values at the fade location using:

before forward processing - initialize

K₁ =K₁₋₋ fade-K₁₋₋ slope·fade₋₋ location

K₂ =K₂₋₋ fade-K₂₋₋ slope·fade₋₋ location

before reverse processing - initialize

K₁ =K₁₋₋ fade-K₁₋₋ slope·(177-fade₋₋ location)

K₂ =K₂₋₋ fade-K₂₋₋ slope·(177-fade₋₋ location)

during processing - as each symbol is processed

K₁ =K₁ +K₁₋₋ slope

K₂ =K₂ +K₂₋₋ slope,

where K₁₋₋ jade is the real value of K₁ at the symbol with the maximumestimated fade depth, K₂₋₋ is the real value of K₂ at the symbol withthe maximum estimated fade depth, K₁₋₋ slope is the real increment in K₁applied during processing of each symbol, K₂₋₋ slope is the realincrement in K₂ applied during processing of each symbol, and fade₋₋location is the symbol number at the maximum estimated fade depth, andlast₋₋ location is the symbol number of the final symbol.

Estimation of the location of the power fade entails use of the receivedsymbols from the matched filter 27, and the settings on the AGC circuit35 that were active during reception of those symbols. As the responseof the amplifier 22 to the AGC circuit settings is effectivelyinstantaneous, the primary delays in utilizing this information arise inthe matched filter 27. This filter 27 is a linear phase filter (constantdelay), so the available input information can be easily transformedinto an accurate estimate of the envelope power. This envelope isaveraged by a rectangular HR filter over about ten (10) symbol times,with very good performance.

After completion of acquisition, the carder frequency offset should beless than 200 Hz. To operate without impairment, this offset should beon the order of 20 Hz or less. Thus, estimation of and correction forcarder offset must continue after acquisition. The method employedutilizes the fact that when frequency offset occurs, the taps of thechannel impulse response will rotate consistently at a rate proportionalto the offset. Changes in tap phases over fixed periods, therefore,provide an observable characteristic to apply to frequency control. Notethat random phase changes occur in addition to these consistent rates ofchange, so filtering is used to extract the frequency offset. Inpractice, offsets of around 1000 Hz can be resolved although the maximumexpected offset after acquisition is 200 Hz. The approach used is asfollows:

1. During the reception of each burst, the half of that burst that doesnot include the deepest fade is selected for tracking. This scheme isaimed at avoidance of the very high rates of change in phase thattypically accompany transitions through low signal amplitudes.

2. Two samples of each of the two estimated channel impulse responsetaps are recorded: just after the preamble (or leading into thepostamble if the fade occurred during the first half of the slot), and20 symbols later (or 20 symbols earlier). At a symbol rate of 24300symbols per second, a 100 Hz offset would result in an average rotationof 29.6 degrees during the 20 symbol period. For any rotation in excessof 180 degrees, the observed rotation would be less than 180 degrees butin the opposite direction. This aliasing could impact performance forfrequency offsets above about 300 Hz. In typical operation, however, thedetriment to performance resulting from such aliasing has provedminimal, due to the anti-aliasing filtering inherent in the tracking.The selection of a sampling window of 20 symbols was based on concernabout this aliasing. Otherwise, a longer window would improve noiseimmunity.

3. From information determined during the bit timing fine tuning, thedominant tap is selected. Using the recorded settings for this tap, aphase change is calculated, yielding an estimate of the frequencyoffset.

4. These estimates are then filtered over many bursts to reduce the"noise" that arises primarily due to the random (zero mean) presence ofDoppler offsets and Gaussian noise. The filter output provides anestimate of the carrier offset and can be used to directly update thefrequency control hardware. The offset is given by:

f₋₋ offset₋₋ estimate_(k+1) =(1-K_(fo))f₋₋ offset₋₋ estimate_(k+) K_(fo)freq₋₋ observed, where freq₋₋ observed is derived from the observedphase change, the constant K_(fo) controls the convergence rate of theestimation process, f₋₋ offset₋₋ estimate_(k) is the estimated frequencyoffset at frame "k", and K_(fo) is a constant controlling theconvergence rate of the frequency tracking. If f₋₋ offset₋₋ estimatereaches half the resolution of the frequency source, then a step infrequency is applied, e.g., if the resolution is 20 Hz and f₋₋ offset₋₋estimate exceeds 10 Hz, then a 20 Hz change in reference is applied. Atthe same time f₋₋ offset₋₋ estimate is reinitialized.

Referring to FIG. 5, it illustrates a flow diagram showing theprocessing performed by the equalizer 20 to implement carrier frequencyoffset compensation. Utilizing an already located fade, a decision (box100) is made as to whether to use the first or second half of thereceived slot for frequency offset estimation. Based on this decision,samples are taken twenty symbols apart in the appropriate half of theslot (boxes 101,102). For the selected case, individual taps arecompared and the larger is chosen (decisions 103, 104). The phases ofthe chosen tap at the selected two times are then subtracted (boxes105-108) to produce "freq₋₋ observed", a noisy estimate of the offset.This is filtered (box 109) to generate an accurate estimate of theoffset. If an adjustment in setting of the frequency control wouldreduce this offset, then a decision is made to do so (decision 110); andthe decision is then implemented (box 111).

The equalizer is reasonably insensitive to errors in bit timing.However, for the following reasons, symbol timing adjustments continueduring equalizer operation. The initial estimate produced by acquisitionmay differ sufficiently from optimal timing so that performance wouldbenefit from adjustment. The transmit and receive symbol timing clocksmay differ by about 5 ppm, resulting in drift of about 0.1 μS per frame(or a symbol every 8 seconds). This drift must be compensated for. Inpractice, individual independently-delayed signal paths will randomlyrise and diminish in average strength, resulting in situations thatwould be best catered for by different symbol timing. Optimal symboltiming depends on an ability to track these changing situations.

The operation of the symbol timing control is as follows. The approachhas similarities to the early-late gating schemes frequently employed indirect-sequence spread spectrum receivers. As each burst is received, ameasure of the error between the expected preamble and the actualreceived preamble is generated. In addition, in alternating frames,similar measures are made on time advanced and retarded versions of thesame input samples. If no timing adjustment is necessary, the errorgenerated with the existing timing should be less (on average) thaneither of the others. Adjustments are made when this is not the case orthere is a consistent disparity between the advanced and retarded errorestimates. This process is simply a search for bit timing that minimizesthe error statistic, as illustrated in FIG. 4. The control loop usedincludes an estimator of any consistent change in timing, correspondingto drift with respect to the transmitter. Drift in the order of 10 ppmcan be compensated for by this loop.

This search for a minimum may be hampered by the possible presence of alocal (non-global) minimum. In fact, for this statistic the presence oftwo minima is common (corresponding to the two taps implicit in theequalizer structure - see FIG. 1 ). The approach taken to resolve thisconflict is as follows. The more advanced minimum is presumed to be thepreferred sampling time. Multiple minima typically arise when there is asmall level of delay spread, i.e., less than about 10 μS. Under suchconditions the ratio of magnitudes of the estimated paths in the(symbol-spaced) channel impulse response differs significantly in theregion of the more advanced minimum from that in the more retarded case.Thus, the ratio of tap magnitudes provides a statistic from which toconclude the appropriateness of a selected minimum.

With reference to FIGS. 6A and 6B they show flow diagrams illustratingthe processing performed by the equalizer 20 to implement bit timingcontrol. Inputs (box 80) include the on-time and time-offset samples (z(n) and z offset(n)), and a flag to indicate the direction of the timeoffset. The on-time samples are fed into the equalizer 20 just as theyare during normal training 83. Similarly, the time offset samples arefed to the equalizer 20 (box 84). In both cases, the branch metrics (onthe known correct paths) are accumulated over the latter symbols toprovide measures (ERROR_(cum) and ERROR OFFSET_(cum)) of the degree towhich the samples match expectations.

In a separate process the magnitudes of each of two taps estimated asthe channel impulse response at the end of the training process arecalculated (box 85). Averaging the ratio of these taps over a number offrames (boxes 86-89) permits a judgement to be made as to whether thebit timing has selected an inappropriate local minimum. If a threshold(box 90) is reached, then bit timing will be advanced by a full symboltime (box 91 ). Taking account of the relative time at which sampleswere taken (box 92), the ERROR_(cum) and ERROR OFFSET_(cum) measures arecombined to generate a noisy estimate of an appropriate timingadjustment (boxes 93, 94). This estimate is then filtered (box 95) togenerate an actual timing offset adjustment. To compensate forconsistent drift, an additional term "drift₋₋ est" monitors andcompensates for this effect.

Thus there has been described a maximum likelihood sequence estimationbased equalization method for use in mobile digital cellular receivers.It is to be understood that the above-described embodiments are merelyillustrative of some of the many specific embodiments which representapplications of the principles of the present invention. Clearly,numerous and other arrangements can be readily devised by those skilledin the art without departing from the scope of the invention.

What is claimed is:
 1. In a digital communications receiver in whichsignals received sequentially within time slots of a transmissionchannel, having a carrier and an impulse response are sampled, a methodfor adjusting timing of the signal sampling and for tracking offset ofthe transmission channel carrier comprising the steps of:storing samplesof the sequentially received signals in a time sequential manner;generating transmission channel impulse response estimates associatedwith the stored samples, the estimates having associated taps;generating from the samples an error measurement comprising a firstmeasure of the degree to which a channel impulse response estimatematches the actual channel impulse response; determining a sample timesetting that minimizes the above error measurement and adjusting thetiming of the signal sampling in accordance with the newly determinedsample time setting; recording at least two samples of at least one tapof a channel impulse response estimate at at least one selected locationwithin a time slot; estimating the location within the time slot atwhich any fade in power received over the channel is a maximum; the timeslot having a preamble and a postamble; during the reception of eachtime slot, selecting the half of the time slot that does not include theestimated location of maximum power fade; recording two samples of eachof two taps of the channel impulse response estimates, the first tapbeing just after the preamble or in the postamble, depending upon theestimated location of maximum power fade, and the second tap being apredetermined number of symbols later or earlier, depending upon theestimated location of maximum power fade; selecting a dominant tap;using recorded settings for the selected dominant tap, calculating aphase change, yielding an estimate of the frequency offset; filteringthe estimate of the frequency offset over many time slots to generatingan estimate of the offset of the transmission channel carrier adaptedfor use as a feedback signal to adjust carrier tracking; and adjusting acontrollable frequency source to track the offset of the transmissionchannel carrier using the feedback signal.
 2. The method of claim 1wherein the step of determining a sample time setting comprisesgenerating a plurality of measures similar to the first measure of thedegree to which the estimated channel impulse response matches theactual channel impulse response utilizing simultaneously recordedsamples having different time offsets wherein at least one sample usedfor each of the plurality of measures is advanced in time and one sampleused for each of the plurality of measures is retarded in time relativeto the samples used for generating the first measure, and furthercomprising filtering the plurality of measures.
 3. The method of claim 1wherein the step of generating channel impulse response estimatescomprises using variable tap coefficients that are determined byestimating tap settings for the estimated channel impulse response byminimizing the square of the difference between actual received samplesand those synthesized by passing known transmitted signals through theestimated channel, and wherein the processing is done in an iterativemanner by combining previous estimates of channel impulse response andnew estimates thereof based on recent estimates, and by varying theratio of the contributions from the previous and new estimates as afunction of location within the time slot.
 4. The method of claim 3wherein the values of coefficients K₁ and K₂ are varied as a function ofsymbol location (k) by solving the following equations for (k+1) and C₁(k+1): ##EQU3## where C₀ (k) and C₁ k) are complex values of estimatedchannel impulse response taps, C_(s0) (k) and C_(n) (k) are complexintermediate values related to the estimated channel impulse responsetaps, permitting second order operation, K₁ and K₂ are the real gainvalues controlling the tracking rate of a channel impulse responseestimation process, z(k) are complex symbol spaced sampled outputs ofthe receiver matched filter, and a(k) are complex estimated or knownvalues of transmitted symbols.
 5. The method of claim 1 wherein the stepof combining the frequency offset estimates comprises applying f₋₋offset₋₋ estimate_(k+1) =(1-K_(fo)) f₋₋ estimate_(k) +K_(fo) freq₋₋observed, where freq₋₋ observed is derived from the observed phasechange, the constant K_(fo) controls the convergence rate of theestimation process, f₋₋ offset₋₋ estimate_(k) is the estimated frequencyoffset at time slot "k", and K_(of) is a constant controlling theconvergence rate of the frequency tracking.
 6. A digital communicationsreceiver in which signals received sequentially within time slots of atransmission channel having a carrier and an impulse response aresampled comprising:a memory for sorting samples of the sequentiallyreceived signals in a time sequential manner, each sample having anassociated phase; a channel impulse response estimate generator coupledto the memory; means coupled to the memory for generating from thestored samples an error measurement comprising a first measure of thedegree to which a channel impulse response estimate matches the actualchannel impulse response; a second memory for recording at least twosamples of at least one tap of a channel impulse response estimate at atleast one selected location within a time slot; a fade locator forestimating the location within the time slot at which any fade in powerreceived over the channel is a maximum the time slot having a preambleand a postamble; means for selecting the half of the time slot that doesnot include the estimated location of maximum power fade wherein thesecond memory records two samples of each of two taps of the channelimpulse response estimates, the first tap being just after the preambleor in the postamble, depending upon the estimated location of maximumpower fade, and the second tap being a predetermined number of symbolslater or earlier, depending upon the estimated location of maximum powerfade; a dominant tap selector; means for using recorded settings fromthe second memory for the selected dominant tap, calculating a phasechange, yielding an estimate of the frequency offset; and a frequencyoffset filter for filtering the estimate of the frequency offset overmany time slots to generate an estimate of the offset of thetransmission channel carrier adapted for use as a feedback signal toadjust carrier tracking; and a controllable frequency source adjustableto track the offset of the transmission channel carrier using thefeedback signal.
 7. The receiver of claim 6 wherein the means forgenerating an error measurement comprises means for generating aplurality of measures similar to the first measure of the degree towhich the estimated channel impulse response matches the actual channelimpulse response utilizing simultaneously recorded samples havingdifferent time offsets from the second memory wherein at least onesample used for each of the plurality of measures is advanced in timeand one sample used for each of the plurality of measures is retarded intime relative to the samples used for generating the first measure; andwherein the means for determining comprises a filter for filtering theplurality of measures.
 8. The receiver of claim 6 wherein the channelimpulse response estimate generator comprises means for using variabletap coefficients that are determined estimates by estimating tapsettings for the channel impulse response by minimizing the square ofthe difference between actual received samples and those synthesized bypassing known transmitted signals through the estimated channel, andwherein the processing is done in an iterative manner by combiningprevious estimates of channel impulse response and new estimates thereofbased on recent estimates, and by varying the ratio of the contributionsfrom the previous and new estimates as a function of location within thetime slot.
 9. The method of claim 8 wherein the values of coefficientsK₁ and K₂ are varied as a function of symbol location (k) by solving thefollowing equations for (k+1) and C₁ (k+1): ##EQU4## where C₀ (k) and C₁(k) are complex values of estimated channel impulse response taps,C_(s0) (k) and C_(n) (k) are complex intermediate values related to theestimated channel impulse response taps, permitting second orderoperation, K₁ and K₂ are the real gain values controlling the trackingrate of the channel impulse response estimation process, z(k) arecomplex symbol spaced sampled outputs of a receiver matched filter, anda(k) are complex estimated or known values of transmitted symbols. 10.In a digital communications receiver in which signals receivedsequentially within time slots of a transmission channel, having acarrier and an impulse response are sampled, a method for adjustingtiming of the signal sampling and for tracking offset of thetransmission channel carrier comprising the steps of:storing samples ofthe sequentially received signals in a time sequential manner;generating transmission channel impulse response estimates associatedwith the stored samples, the estimates having associated taps; usingvariable tap coefficients that are determined by estimating tap settingsfor the estimated channel impulse response by minimizing the square ofthe difference between actual received samples and those synthesized bypassing known transmitted signals through the estimated channel, theprocessing being done in an iterative manner by combining previousestimates of channel impulse response and new estimates thereof based onrecent estimates, and by varying the ratio of the contributions from theprevious and new estimates as a function of location within the timeslot, the ratio being performed by varying the values of coefficients K₁and K₂ as a function of symbol location (k) by solving the followingequations for (k+1) and C₁ (k+1): ##EQU5## where C₀ (k) and C₁ (k) arecomplex values of estimated channel impulse response taps, C_(s0) (k)and C_(n) (k) are complex intermediate values related to the estimatedchannel impulse response taps, permitting second order operation, K₁ andK₂ are the real gain values controlling the tracking rate of the channelimpulse response estimation process, z(k) are complex symbol spacedsampled outputs of a receiver matched filter, and a(k) are complexestimated or known values of transmitted symbols; generating from thesamples an error measurement comprising a first measure of the degree towhich a channel impulse response estimate matches the actual channelimpulse response; determining a sample time setting that minimizes theabove error measurement and adjusting the timing of the signal samplingin accordance with the newly determined sample time setting; recordingat least two samples of at least one tap of a channel impulse responseestimate at least one selected location within a time slot; generating afrequency offset estimate from the phase difference in the time slotbetween the at least two samples using the recorded tap samples; andadjusting a controllable frequency source to track the offset of thetransmission channel carrier in accordance with the frequency offsetestimate.
 11. The method of claim 10 wherein the step of generating afrequency offset estimate comprises:generating frequency offsetestimates for each of a plurality of time slots from the phasedifference in each time slot between at least two samples of a singletap of a channel impulse response estimate; and combining this pluralityof frequency offset estimates to generate a precise frequency offsetestimate; and wherein the step of adjusting comprises adjusting tocompensate for the precise frequency offset estimate.
 12. The method ofclaim 11 wherein the step of generating frequency offset estimatescomprises:selecting a dominant tap; using recorded tap samples from thedominant tap; and calculating a phase change using the recorded tapsamples, yielding an estimate of the frequency offset.
 13. The method ofclaim 11 wherein each time slot includes a preamble and a postamble andwherein the step of generating frequency offset estimatescomprises:estimating the location within the time slot at which any fadein power received over the channel is a maximum; during the reception ofeach time slot, selecting the half of the time slot that does notinclude the estimated location of maximum power fade; recording twosamples of each of two taps of the channel impulse response estimates,the first tap being just after the preamble or in the postamble,depending upon the estimated location of maximum power fade, and thesecond tap being a predetermined number of symbols later or earlier,depending upon the estimated location of maximum power fade; selecting adominant tap; using recorded settings for the selected dominant tap,calculating a phase change, yielding an estimate of the frequencyoffset; and filtering the estimate of the frequency offset over manytime slots to generate an estimate of the carder offset adapted for useas a feedback signal to adjust carrier tracking.
 14. The method of claim11 wherein the step of combining the frequency offset estimatescomprises applying f₋₋ offset₋₋ estimate_(k+1) =(1-K_(fo))f₋₋ estimate_(k) +K_(fo) freq₋₋ observed, where freq₋₋ observed is derived from theobserved phase change, the constant K_(of) controls the convergence rateof the estimation process, f₋₋ offset₋₋ estimate_(k) is the estimatedfrequency offset at time slot "k", and K_(of) is a constant controllingthe convergence rate of the frequency tracking.
 15. The method of claim10 further comprising:estimating the location within the time slots atwhich a data value estimate error is most probable; and processing thestored samples, starting with the first received sample in a time slotand proceeding in a forward direction with respect to the time sequencein which the samples were stored, beyond the estimated most probableerror location, using a preselected maximum likelihood sequenceestimation procedure to generate estimates of data values transmitted inthe time slots.
 16. The method of claim 15 further comprising the stepsof processing the stored samples, starting with the final receivedsample in the time slot and proceeding in a reverse direction withrespect to the time sequence in which the samples were stored, beyondthe estimated most probable error location, using the maximum likelihoodsequence estimation procedure to generate estimates of the transmitteddata values; andprocessing the estimates of the preceding processingsteps to generate enhanced estimates of the values of the transmitteddata sequence.
 17. The method claim 16 wherein the step of processingthe estimates comprises applying the transmission channel impulseresponse estimates to the maximum likelihood sequence estimationprocedures for generating enhanced estimates.
 18. A digitalcommunications receiver in which signals received sequentially withintime slots of a transmission channel having a carrier and an impulseresponse are sampled comprising:a memory for storing samples of thesequentially received signals in a time sequential manner, each samplehaving an associated phase; a channel impulse response estimategenerator coupled to the memory, comprising means for using variable tapcoefficients that are determined by estimating tap settings for theestimated channel impulse response estimates by minimizing the square ofthe difference between actual received samples and those synthesized bypassing known transmitted signals through the estimated channel, whereinthe processing is done in an iterative manner by combining previousestimates of channel impulse response and new estimates thereof based onrecent estimates, and by varying the ratio of the contributions from theprevious and new estimates as a function of location within the timeslot by varying the values of coefficients K₁ and K₂ as a function ofsymbol location (k) by solving the following equations for (k+1) and C₁(k+1): ##EQU6## where C₀ (k) and C₁ (k) are complex values of estimatedchannel impulse response taps, C_(s0) (k) and C_(n) (k) are complexintermediate values related to the estimated channel impulse responsetaps, permitting second order operation, K₁ and K₂ are the real gainvalues controlling the tracking rate of the channel impulse responseestimation process, z(k) are complex symbol spaced sampled outputs of areceiver matched filter, and a(k) are complex estimated or known valuesof transmitted symbols; means coupled to the memory for generating fromthe stored samples an error measurement comprising a first measure ofthe degree to which a channel impulse response estimate matches theactual channel impulse response; means for determining a sample timesetting that minimizes the above error measurement and for adjusting thetiming of the signal sampling in accordance with the newly determinedsample time setting; a second memory for recording at least two samplesof at least one tap of a channel impulse response estimate at least oneselected location within a time slot; means for generating a frequencyoffset estimate from the phase difference in the time slot between theat least two samples using the recorded tap samples; and a controllablefrequency source adjustable to track the offset of the transmissionchannel carrier in accordance with the frequency offset estimate. 19.The receiver of claim 18 further comprising:means coupled to the channelimpulse response estimator for generating frequency offset estimates foreach of a plurality of time slots from a phase difference in each timeslot between at least two samples of a single tap of a channel impulseresponse estimate; and a filter for combining this plurality offrequency offset estimates to generate a precise frequency offsetestimate, and wherein the frequency source is adjustable to compensatefor the precise frequency offset estimate.
 20. The receiver of claim 19wherein each sample recorded in the second memory includes a preambleand a postamble and wherein the means for generating frequency offsetestimates comprises:a fade locator for estimating the location withinthe time slot at which any fade in power received over the channel is amaximum; means for selecting the half of the time slot that does notinclude the estimated location of maximum power fade wherein the secondmemory records two samples of each of two taps of the channel impulseresponse estimates, the first tap being just after the preamble or inthe postamble, depending upon the estimated location of maximum powerfade, and the second tap being a predetermined number of symbols lateror earlier, depending upon the estimated location of maximum power fade;a dominant tap selector; means for using recorded settings from thesecond memory for the selected dominant tap, calculating a phase change,yielding an estimate of the frequency offset; and a frequency offsetfilter for filtering the estimate of the frequency offset over many timeslots to generate an estimate of the carrier offset adapted for use as afeedback signal to adjust carrier tracking.
 21. In a digitalcommunications receiver in which signals received sequentially withintime slots of a transmission channel, having a carrier and an impulseresponse are sampled, a method for adjusting timing of the signalsampling and for tracking offset of the transmission channel carriercomprising the steps of:storing samples of the sequentially receivedsignals in a time sequential manner; generating transmission channelimpulse response estimates associated with the stored samples, theestimates having associated taps; generating from the samples an errormeasurement comprising a first measure of the degree to which a channelimpulse response estimate matches the actual channel impulse response;determining a sample time setting that minimizes the above errormeasurement and adjusting the timing of the signal sampling inaccordance with the newly determined sample time setting; recording atleast two samples of at least one tap of a channel impulse responseestimate at at least one selected location within a time slot;generating frequency offset estimates for each of a plurality of timeslots from the phase difference in each time slot between at least twosamples of a single tap of a channel impulse response estimate;combining this plurality of frequency offset estimates to generate aprecise frequency offset estimate by applying f₋₋ offset₋₋estimate_(k+1) =(1-K_(fo)) f₋₋ estimate_(k) +K_(fo) freq₋₋ observed,where freq₋₋ observed is derived from the observed phase change, theconstant K_(fo) controls the convergence rate of the estimation process,f₋₋ offset₋₋ estimate_(k) is the estimated frequency offset at time slot"k", and K_(of) is a constant controlling the convergence rate of thefrequency tracking; and adjusting a controllable frequency source totrack the offset of transmission channel carrier in accordance with theprecise frequency offset estimate.
 22. The method of claim 21 whereinthe step of determining a sample time setting comprises generating aplurality of measures similar to the first measure of the degree towhich the estimated channel impulse response matches the actual channelimpulse response utilizing simultaneously recorded samples havingdifferent time offsets wherein at least one sample used for each of theplurality of measures is advanced in time and one sample used for eachof the plurality of measures is retarded in time relative to the samplesused for generating the first measure, and further comprising filteringthe plurality of measures.
 23. The method of claim 21 wherein the stepof generating frequency offset estimates comprises:selecting a dominanttap; using recorded tap samples from the dominant tap; and calculating aphase change using the recorded tap samples, yielding an estimate of thefrequency offset.
 24. The method of claim 21 wherein each time slotincludes a preamble and a postamble and wherein the step of generatingfrequency offset estimates comprises:estimating the location within thetime slot at which any fade in power received over the channel is amaximum; during the reception of each time slot, selecting the half ofthe time slot that does not include the estimated location of maximumpower fade; recording two samples of each of two taps of the channelimpulse response estimates, the first tap being just after the preambleor in the postamble, depending upon the estimated location of maximumpower fade, and the second tap being a predetermined number of symbolslater or earlier, depending upon the estimated location of maximum powerfade; selecting a dominant tap; using recorded settings for the selecteddominant tap, calculating a phase change, yielding an estimate of thefrequency offset; and filtering the estimate of the frequency offsetover many time slots to generating an estimate of the carrier offsetadapted for use as a feedback signal to adjust carrier tracking.
 25. Themethod of claim 21 wherein the step of generating channel impulseresponse estimates comprises using variable tap coefficients that aredetermined by estimating tap settings for the estimated channel impulseresponse by minimizing the square of the difference between actualreceived samples and those synthesized by passing known transmittedsignals through the estimated channel, and wherein the processing isdone in an iterative manner by combining previous estimates of channelimpulse response and new estimates thereof based on recent estimates,and by varying the ratio of the contributions from the previous and newestimates as a function of location within the time slot.
 26. The methodof claim 25 wherein the values of coefficients K₁ and K₂ are varied as afunction of symbol location (k) by solving the following equations for(k+1) and C₁ (k+1): ##EQU7## where C₀ (k) and C₁ (k) are complex valuesof estimated channel impulse response taps, C_(s0) (k) and C_(n) (k) arecomplex intermediate values related to the estimated channel impulseresponse taps, permitting second order operation, K₁ and K₂ are the realgain values controlling the tracking rate of the channel impulseresponse estimation process, z(k) are complex symbol spaced sampledoutputs of a receiver matched filter, and a(k) are complex estimated orknown values of transmitted symbols.
 27. The method of claim 21 furthercomprising:estimating the location within the time slots at which a datavalue estimate error is most probable; and processing the storedsamples, starting with the first received sample in a time slot andproceeding in a forward direction with respect to the time sequence inwhich the samples were stored, beyond the estimated most probable errorlocation, using a preselected maximum likelihood sequence estimationprocedure to generate estimates of data values transmitted in the timeslots.
 28. The method of claim 27 further comprising the steps ofprocessing the stored samples, starting with the final received samplein the time slot and proceeding in a reverse direction with respect tothe time sequence in which the samples were stored, beyond the estimatedmost probable error location, using the maximum likelihood sequenceestimation procedure to generate estimates of the transmitted datavalues; andprocessing the estimates of the preceding processing steps togenerate enhanced estimates of the values of the transmitted datasequence.
 29. The method claim 28 wherein the step of processing theestimates comprises applying the transmission channel impulse responseestimates to the maximum likelihood sequence estimation procedures forgenerating enhanced estimates.